Current-Fed Isolated DC/DC Converter for Future Aerospace Microgrids

High-performance power conversion equipment is currently gaining an increasing interest for aircraft applications. In particular, isolated bidirectional dc/dc converters are often proposed for modern aircraft distribution systems. A current-fed isolated dc/dc converter, named active-clamp active-bridge topology, is identified as the most promising for the proposed application, interfacing a 270-V dc network with a 28-V dc network. A comparison between the selected topology and the well-known dual-active-bridge topology has been carried out and an experimental prototype has been manufactured for the selected conversion architecture. Simulation and experimental results are provided in order to validate the tradeoff and the design of the proposed converter.


I. INTRODUCTION
In recent years, the More Electric Aircraft (MEA) concept has gained an increased importance [1]. In fact, electrical systems are replacing hydraulic, mechanical, or pneumatic power sources in a wide range of aerospace applications [2]. This increase in electrical energy demand has led to a rapid technology development, particularly in power electronics [3]. Electrical systems are now considered for aircraft actuation systems, wing ice protection systems, environmental control systems, and fuel pumping. These novel electrical systems aim to increase future aircraft efficiency, thus reducing the environmental impact of such systems and their maintenance cost. However, due to the electrical system complexity, future aircraft will face similar issues to the one found in ground based microgrids [1]- [3].In such scenario, several structures for future aerospace microgrids have been proposed. Both AC and DC grids are investigated showing a trend in increasing the voltage level in such grids [4], [5]. Focusing on High Voltage Direct Current (HVDC) aerospace microgrids, Fig. 1 shows a typical grid structure which comprises a Low Voltage Direct Current (LVDC) network and a HVDC network. On the LVDC network typically Low Voltage (LV) emergency batteries and other energy storage elements, such as Electro-Mechanical Batteries (EMB) are connected together with the Electrical Power Unit (EPU) which supplies avionics circuitry [6]. On the HVDC network, High Voltage (HV) generators [7], [8] provide the required voltage and power, together with HV batteries [6] and other EPU [9], which supplies electric actuators, de-icing systems and other MEA equipment [10]- [12]. Moreover, Starter Generator (SG) systems [13], [14], together with their Electrical Control Unit (ECU) are able to provide the required power to start the aircraft engines, and generate additional power during flight, on the HVDC or LVDC network. As shown in Fig.1, an isolated, bidirectional DC/DC converter has to be included in the system in order to provide an active interface between the two DC networks [15], [16]. The converter must be designed to provide high power density and high efficiency, to be integrated in the aircraft structure with minimum impact on volume, weight and heat management. At the same time, it must provide reliable and flexible operation, also capable to respond to fault events or other abnormal operating conditions with a controlled and predictable behaviour, while minimising the impact on power generation. In particular, the reliability of the DC-DC converter is crucial during "emergency flight" operation, as it will supply the essential equipment to guarantee flight safety. The converter will also operate in a stringent and harsh environment, including limited cooling capabilities; pressure drops, mechanical vibrations and possible EMC or lightning interference, and their operation should have minimum effect on the environment. In order to achieve these challenging requirements, it is clear that cutting-edge technological innovations and innovative design approaches will be required. Several isolated DC/DC converter topologies are proposed in literature. Flyback, Forward, and Forward-Flyback converters are investigated in [17], [18]. These topologies requires a minimal number of active devices, however, their low efficiency, compared with other converter structures limits their use in high power applications. Resonant converters, such as isolated LLC resonant DC/DC converter and Series Resonant (SR) converter [19] can also be considered for this application. They take advantage of resonant networks in order to achieve Zero Voltage Switching (ZVS) or Zero Current Switching (ZCS) commutations, providing highly efficient power conversion. However, additional passive components are needed and control becomes challenging when a wide operating area is required by the application. On the other hand, Dual-Active-Bridge (DAB) [20] and Three-Phase Dual-Active-Bridge [21] present lower loss switching loss due to their natural capability of achieving ZVS commutation when the devices are turned on, in a wide operating range and without drastically increase the control system complexity. In unidirectional applications, Synchronous rectification also represents an high efficiency alternative to DAB [22]. However, it is not applicable to this work since bidirectional power flow is required. In alternative to DAB a current fed solution has been proposed in previous work, named Active Bridge Active Clamp (ABAC) converter [23]. In fact, the ABAC converter represent a promising solution for MEA applications, due to its inherent current control capabilities. In the following sections, the ABAC converter is introduced and compared with the DAB in terms of design, operating capabilities and weight and volume considerations. Simulations results are provided in several operating conditions in order to assess the benefits of ABAC converter. Experimental results are also provided on a 10kW prototype.

II. THE DUAL ACTIVE BRIDGE CONVERTER
The DAB converter [24], [25], shown in Fig. 2 is a wellknown topology, which provides efficient DC/DC conversion as well as galvanic insulation. As shown in Fig. 2, the DAB consists of two H-Bridge linked on their AC sides with a series inductor and an HF transformer. Conventional Single Phase Shift (SPS) modulation scheme [26] is adopted. With SPS the duty cycle of each bridge/arm is kept constant at 50% of the sampling period, while the phase shift φ between the transformer primary and secondary voltage waveform is used to control the power transfer. Theoretical operation waveforms for such topology is drawn in Fig. 3, where Vp and Vs are transformer primary and secondary voltages, which operate at 50% duty cycle in SPS modulation, ip is transformer primary current. iLV and ILV are the currents flowing before and after being filtered by the output capacitor CLV. It is noted that the output current iLV has considerably large ripple which needs a significantly large capacitive filter. The power transfer inductance Lex on the primary side is designed in order to achieve the required power rating for the converter [20], where, N is transformer turn ratio, VHV and VLV are high DC bus voltage and low DC bus voltage respectively, fs is switching frequency and Pmax is the rated power. In order to design the filter capacitors, Ip1 and Ip2 in Fig. 3 Referring to Fig. 3, the time intervals Δ1 and Δ2 can be calculated using the following equations: Therefore, the peak-to-peak output voltage ripple rV can be obtained based on following equation: where, The output capacitor CLV can then be derived from (6) as: A similar approach can be used to calculate the value of CHV. In order to analyse the ZVS operating area for the DAB converter the voltage transfer ratio M is defined: In buck operations, i.e. when the power is transferred from primary to secondary converter side, the secondary phase shift ϕ is positive and ZVS is achieved when Ip1>0 and Ip2>0.Thus, from (2), (3) and (10) the following conditions can be obtained: Similarly in boost operations, i.e. when the power is transferred from secondary to primary converter side, the secondary phase shift ϕ is negative and ZVS is achieved when Ip1<0 and Ip2<0, obtaining the following constraint: In Fig. 4, the boundaries (11) and (12)   The control scheme adopted with DAB converter is shown in Fig. 5 for both Buck mode, where the power flows from the HV network to a LV load, and Boost mode, where the power flows from the LV network to a HV load. Clearly only voltage control is feasible with DAB converter without adding additional inductors in the circuit. In terms of modulation, conventional Single Phase Shift (SPS) modulation is implemented. In this case the duty cycle of the HV and LV bridge, dHV and dLV respectively, are kept fixed at 50% of the sampling interval Ts. Implementation of SPS is straightforward but presents numerous disadvantages, such as large back-andforth power [27], hard switching [20], [24], dead band effect [26] under light load non-nominal voltages operation condition. In order to overcome these problems, advanced modulations has been proposed in literature, such as Extended Phase Shift (EPS) [28], Dual Phase Shift (DPS) [29], Triple Phase Shift (TPS) [30], Triangular current Manipulation (TRM) and Trapezoidal current Manipulation (TZM) [31] .

III. THE ACTIVE BRIDGE ACTIVE CLAMP CONVERTER
The ABAC converter, shown in Fig. 6, features a similar operating behaviour to the DAB. The main difference is that it provides a current-fed LV stage, taking advantage of external output inductors. The power transfer inductor Lex transfers power from primary to the clamping circuits, while the two output inductors, L1 and L2, serves as buffers to transfer energy from the clamping circuits to the LV output. When using SPS modulation, the switches are all operating at 50% duty, and the two clamping circuits are complementarily switched in order to produce a square waveform on the transformer primary. The design of the input capacitor CHV is identical to the one in DAB. Similarly, the power transfer inductor Lex can be designed in order to obtain the required power transfer: where N is the transformer turn ratio for the ABAC converter, and the voltage on the clamping capacitors vC1 and vC2 have an average value of twice the amplitude of VLV when the converter is operating with SPS modulation and, thus, 50% of duty cycle. The main advantage of ABAC converter, with respect to DAB, is the capability of obtaining interleaved LV inductor currents iLV1 and iLV2, resulting, ideally, in a purely DC current ILV. Thus, there is no need of capacitive filtering on the LV side of the converter. However, a filter capacitance is used to suppress the current ripple related to converter asymmetries and load harmonics. Assuming a desired current peak-to-peak ripple before the interleaved point is rI, the constraint for the inductors L1 and L2 can be defined as: Regarding ZVS turn-on operating region for the ABAC converter, the primary side boundary condition is derived as for DAB, considering that the clamp capacitor voltages vC1 and vC2 are approximately equal to twice the value of VLV As for the DAB converter, the following soft switching constraint is obtained for the primary converter side in equations (16) and (17), for buck and boost operations, respectively.
On the other hand, ZVS constraint for secondary switches turn-on is defined referring to Fig. 7, where the hard turn-off of the switches already took place and, after the dead-time, the complimentary switch on each clamp circuits is turning on. It can be noted that in the case buck operations T5 and T8 are soft switched during turn-on only if the secondary current is is bigger than the respective inductor current. Vice-versa, in the case of boost operations, T6 and T7 are soft switched during turn-on only if the secondary current is is bigger than the respective inductor current, after inverting its sign.  According to this analysis soft-switching conditions can be rewritten as in (18) Equations (18) to (22) lead to complex expressions that can be simplified if a negligible current ripple rI is considered for the output inductors, resulting in the soft-switching region expressed by equations (23) and (24) for buck and boost operations, respectively.
According to the condition in (23) and (24), where rI = 0 is assumed, the ZVS region for ABAC converter is shown in Fig.  9 for various values of N. The results in Fig. 9 represent an ideal case where infinite output inductance is considered and, in terms of soft switching capabilities, represents the worst scenario. When a finite value of output inductance is considered, the increased value of rI widen the soft switching are by adding an offset term to the boundary conditions, as shown in Fig. 10. In this general case the soft-switching region is expressed by equations (25) and (26) for buck and boost operations, respectively, with Lo = L1 =L2 in Fig. 6 From these equations is clear that lower values of Lo may be desirable. However it should also be considered that lower values of Lo results in a higher current ripple and, thus, higher AC losses in the output inductors.
The control scheme adopted with ABAC converter is shown in Fig. 11 for both Buck mode and Boost mode. Conversely to DAB converter, with ABAC converter the implementation of a current control loop is straight forward by regulating the LV current on the output inductors. In terms of modulation, conventional Single Phase Shift (SPS) modulation is implemented. As for DAB converter, the aforementioned limitations of SPS can be avoided by controlling dLV. In fact in ABAC converter the clamp voltages are directly related to VLV and dLV. By controlling dLV in order to keep the amplitude of Vp and Vs well matched, it is possible to ensure soft switching in a wide input and output voltage operating range [32]. However this will affect the maximum power transfer. The advanced modulation aforementioned for DAB cannot be directly applied to ABAC converter since they will affect the output current interleaving [33]. As an alternative, if a split secondary structure is considered, as shown in Fig. 12 it is possible to restore the applicability of these modulation by independently controlling the phase shift of the two secondaries [34], [35]. The power transfer inductance in this case is moved on the secondary side of the converter in order to decouple the dynamics of the two secondary winding. Its value can be easily calculated from equation (13) as follows: The structure of Fig. 12 is completely equivalent to a DAB converters in terms of modulation and it is considered as a term of comparison between the two topologies.

IV. SIMULATION RESULTS
Simulations are carried out considering the parameters of Table I in both Buck (when power is transferred from an HV source to a LV load) and Boost (when power is transferred from an LV source to a HV load) modes.  Fig. 13 show steady state operations in Buck mode, when the DAB and ABAC converters are providing 8.4kW on a LV resistive load. Results show similar input/output performance of the two topologies under investigation. As it can be noted, the DAB presents a considerable current ripple on the LV output, which has to be filtered through the LV capacitor, while ABAC present a continuous LV output current with minimum ripple. Fig. 14 show steady state operations in Boost mode, when the DAB and ABAC converters are providing 6kW on a LV resistive load with the same HV current ripple considerations already discussed for buck mode. Fig. 15 shows the behaviour of the two converters when a short circuit is applied on the LV bus, with a short circuit resistance of 78.4mΩ, at time 0.1s. In this case the DAB converter presents high current ripple on the LV output with a peak value of 950A which is equal to approximately three times the converter rated current. For this reason additional inductors may be required to actively control the current during faults. On the other hand ABAC converter inherently allows current control and present a continuous LV output current, saturated by control at 400A which presents minimal overshoot. In fact the ABAC converter inherently allow current control, i.e. to modify the control output in order to obtain the desired current, which is limited to 400A. When this value is reached, the control output is no longer modified. An efficiency comparison is also shown in Fig. 15. The power electronics devices has been selected accordingly with the required voltage and current rating and paralleled in order to increase the converter current rating or efficiency. In particular, the IPT02N10N3 Silicon Mosfet, rated 100V, 300A has been selected for the converter LV side, while the C2M0025120D Silicon Carbide Mosfet, rated 1.2kV, 90A has been selected for the high voltage side. In order to perform a fair comparison the same number of devices in parallel has been selected for the two converters (one on the HV side and four on the LV side) with the only difference that for the ABAC converter these devices are equally divided between the two secondary circuits. Regarding the passive components, capacitors losses are considered negligible while only the DC losses are considered for the magnetics components. In fact, the AC losses can be minimized by performing an accurate planar magnetics design. The efficiency is estimated through PLECS simulations. The results, shown in Fig, 15, highlight the high efficiency for these two topologies when the soft switching constraints are satisfied. When the soft switching is lost as for example in the 300V/21V curve efficiency drops drastically due to increased switching losses. Advanced modulations can be implemented to increase the soft switching region. It is important to highlight that this efficiency analysis is carried out with the solely scope of comparing the two converter and do not represent an accurate measurement of efficiency, since additional losses in the magnetics are not considered and switches thermal models are obtained using datasheet parameters.
V. WEIGHT AND VOLUME CONSIDERATIONS A first estimation of weight and volume for the three topologies has been calculated by considering off the shelf components and their datasheet information, shown in Table II, combined with the design parameters. The weight and volume estimation does not include heatsink weight and the additional weight and volume of PCB boards, wiring and other components. The design is carried out at three different power rating, 5kW, 10kW and 20kW respectively, in order to identify trends in the components volume and weight. The results in Fig.  16, show that ABAC converter has more weight and volume associated with magnetic components while the DAB converter has an higher capacitor weight and volume. The increased capacitor weight and volume in DAB is mainly related to their demanding current capabilities which requires a combination of film and ceramic capacitors and discourage the use of electrolytic capacitors. On the other hand even if magnetics in the ABAC converter has to handle high currents, their inductance value is relatively low and few turns are required, thus reducing their weight and volume and allowing further optimization of these components. This trend gets more and more evident when increasing the rated power, making the ABAC converter particularly suitable for high power ratings.   VI. EXPERIMENTAL RESULTS A 8.4kW experimental prototype based on the ABAC converter has been manufactured, as shown in Fig. 17. Experimental results are shown in Fig. 18 and Fig.19 when the converter is operating in buck and boost mode, respectively. The results are obtained considering the design parameters of Table I and a power transferred to the load of 3kW and 5kW, in buck and boost mode respectively. Voltages and currents at primary and one secondary of the HF transformer are presented, as well as voltage and current on both converter external interfaces (supply and load). The load voltage is well regulated at 28V, while providing the load current without any noticeable oscillation. On the other hand, looking at the current flowing through one of the four output inductors, it is possible to notice that the triangular ripple, which is cancelled at the load point, matches the value previously obtained in simulations. Experimental results validate the results of simulations except for the inductive voltage drops on the secondary transformer voltage, which are caused from parasitic inductance on the PCB board design. Fig. 18. Experimental results for the proposed ABAC topology in buck mode. vp (red, 500V/div), vs1 (yellow, 100V/div), ip (blue, 50A/div), is1 (green, 100A/div); Voltage and current on HV and LV converter side, Inductor current. Fig. 19. Experimental results for the proposed ABAC topology in boost mode. vp (red, 500V/div), vs1 (yellow, 100V/div), ip (blue, 50A/div), is1 (green, 100A/div); Voltage and current on HV and LV converter side, Inductor current.

VII. CONCLUSIONS
In this work a comparison between DAB converter and ABAC converter for aerospace applications has been carried out. The two topologies are compared in terms efficiency, weight and volume for the specific application of a 270V/28V 10kW bidirectional DC/DC converter. The design procedure has been described and soft switching analysis has been performed for both topologies. Accordingly to these analysis the two converter are evaluated through simulations in term to input and output waveforms, control, and efficiency. Additionally a first estimation of weight and volume for the two topologies when the converter power rating vary from 5kW to 20kW is carried out. Finally, experimental results validate the results obtained by simulations. From the obtained results it can be concluded that the ABAC converter represents a promising alternative to the DAB converter, able to reduce the converter weight and volume at high power ratings and inherently control the LV current whilst maintaining similar efficiency of a DAB converter.